Circuit for reading a charge retention element for a time measurement

ABSTRACT

A method and a circuit for reading an electronic charge retention element for a temporal measurement, of the type including at least one capacitive element whose dielectric exhibits a leakage and a transistor with insulated control terminal for reading the residual charges, the reading circuit including; two parallel branches between two supply terminals, each branch including at least one transistor of a first type and one transistor of a second type, the transistor of the second type of one of the branches consisting of that of the element to be read and the transistor of the second type of the other branch receiving, on its control terminal, a staircase signal, the respective drains of the transistors of the first type being connected to the respective inputs of a comparator whose output provides an indication of the residual voltage in the charge retention element.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to electronic circuits and, morespecifically, to the forming of a circuit enabling controllably storingelectric charges for a time measurement.

2. Discussion of the Related Art

In many applications, it is desired to have data representative of atime elapsed between two events, be it an accurate or approximatemeasurement. An example of application relates to the time management ofrights of access, especially to media.

The obtaining of such data representative of the elapsed timeconventionally requires a time measurement by an electronic circuitpowered, for example by means of a battery, to avoid losing the historyof the data when the circuit is not used.

It would be desirable for a time measurement which operates even whenthe electronic measurement circuit is not powered to be available.

International patent WO-A-03/083769 describes a time-measurement securedtransactional electronic entity in which the time elapsed between twosuccessive transactions is determined by measuring the charge of acapacitive component exhibiting a leakage from its dielectric spacer.The component is charged when the circuit is powered and its residualcharge, after interruption of the power supply, is measured when thecircuit is powered back on. This residual charge is considered asrepresentative of the time elapsed between the two circuit-poweringtimes.

The electronic entity is based on a MOS transistor having its gateconnected to a first electrode of a capacitive component which has itsother electrode grounded along with the transistor source. Thetransistor drain is connected to a power supply voltage by means of acurrent-to-voltage conversion resistor. The voltage measured across theresistor is a function of the drain current in the transistor, and thusof its gate-source voltage, hence of the voltage across the capacitivecomponent. A time interval is initialized by charging the capacitivecomponent by application of an electric power source on its electrodecommon with the transistor gate.

The solution provided by this document has several disadvantages.

First, the measurable time range is limited by the possibilities ofintervention on the capacitive component dielectric.

Then, the charge of the capacitive component generates an electricstress on its dielectric, whereby the measurements drift along time.

Further, the provided structure requires forming of a specificcomponent. In certain applications, it would be desirable to associatethe time measurement element with a memory to condition the access tothe data or programs contained in this memory. The known solution of theabove-mentioned document is not easily compatible with the memorymanufacturing steps.

Further, the interpretation of the residual charge in the capacitivecomponent requires calibration steps to generate charge-to-timeconversion tables.

SUMMARY OF THE INVENTION

At least one embodiment of the present invention aims at overcoming allor part of the disadvantages of known solutions to provide datarepresentative of a time elapsed between two events, without it beingnecessary to permanently power the electronic circuit containing themeans to achieve this.

According to a first aspect, at least one embodiment of the presentinvention aims at an electronic circuit of charge retention for a timemeasurement.

According to a second aspect, at least one embodiment of the presentinvention aims at the forming of such a circuit in a way compatible withtechnologies used to form memory cells.

According to a third aspect, at least one embodiment of the presentinvention aims at the reading from an electronic charge retentioncircuit without the constraint of a table for converting a residualcharge value into a time interval.

According to a fourth aspect, at least one embodiment of the presentinvention aims at a fast programming of an electronic charge retentioncircuit.

To achieve all or part of these objects, as well as others, at least oneembodiment of the present invention provides a circuit for reading froman electronic charge retention element for a time measurement, of thetype comprising at least one capacitive element with a dielectric havinga leakage and a transistor with an isolated control terminal for readingresidual charges, the read circuit comprising:

two parallel branches between two power supply terminals, each branchcomprising at least one transistor of a first type and one transistor ofa second type, the transistor of the second type of one of the branchesbeing formed by that of the element to be read and the transistor of thesecond of the other branch receiving, on its control terminal, a steppedsignal, the respective drains of the transistors of the first type beingconnected to the respective inputs of a comparator having its outputproviding an indication of the residual voltage level in the chargeretention element.

According to an embodiment of the present invention, each branchcomprises a second transistor of the second type having its controlterminals connected at the output of an amplifier of application of areference voltage, said voltage being also applied on a conductionterminal of the transistor of the charge retention element.

According to an embodiment of the present invention, a referenceterminal of the capacitive element of the charge retention element to beread from is connectable to a reference terminal of a second referencecharge retention element.

According to an embodiment of the present invention, said stepped signalis provided by a digital-to-analog converter of non-linear type.

The present invention also provides a method for calibrating a readcircuit, in which a voltage reference value of the digital-to-analogconverter corresponds to Q(r)/(k*C_(T)), where k represents the maximumcount of the converter, Q(r) represents the initial charge of theretention circuit, and C_(T) represents its capacitance.

The present invention also provides a method for calibrating a readcircuit, in which a weighting coefficient of the count provided by theconverter is deduced from an initial measurement from a known timeinterval.

The foregoing and other objects, features, and advantages of the presentinvention will be discussed in detail in the following non-limitingdescription of specific embodiments in connection with the accompanyingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic block diagram illustrating an electronic entityequipped with a charge retention circuit according to an aspect of thepresent invention;

FIG. 2 shows an embodiment of an electronic charge retention circuitaccording to the first aspect of the present invention;

FIG. 3 is a current-vs.-voltage graph illustrating the operation of thecircuit of FIG. 2;

FIG. 4 is a graph illustrating the operation of the circuit of FIG. 2;

FIG. 5 shows a second embodiment of a charge retention circuit accordingto the first aspect of the present invention;

FIG. 6 is a current-vs.-voltage graph illustrating the operation of thecircuit of FIG. 5;

FIG. 7 shows a variation of the circuit of FIG. 5 in an example ofenvironment;

FIGS. 8A, 8B, and 8C respectively are a top view, a cross-section viewalong a first direction and the equivalent electric diagram of anembodiment of an electronic charge retention circuit according to thesecond aspect of the present invention;

FIGS. 9A, 9B, and 9C respectively are a top view, a cross-section viewalong a second direction, and the equivalent electric diagram of a firstelement of the circuit of FIGS. 8A to 8C;

FIGS. 10A, 10B, and 10C respectively are a top view, a cross-sectionview along the second direction and the equivalent electric diagram of asecond element of the circuit of FIGS. 8A to 8C;

FIGS. 11A, 11B, and 11C respectively are a top view, a cross-sectionview along the second direction, and the equivalent electric diagram ofa third element of the circuit of FIGS. 8A to 8C;

FIGS. 12A, 12B, and 12C respectively are a top view, a cross-sectionview along the second direction, and the equivalent electric diagram ofa fourth element of the circuit of FIGS. 8A to 8C;

FIG. 13 shows a first embodiment of a read circuit of an electroniccharge retention circuit according to the third aspect of the presentinvention;

FIG. 14 partially shows a second embodiment of a read circuit of anelectronic charge retention circuit according to the third aspect of thepresent invention;

FIG. 15 shows an example of a non-linear digital-to-analog converterusable in a read circuit according to the third aspect of the presentinvention;

FIGS. 16A and 16B are graphs illustrating an operating mode of a readcircuit according to the third aspect of the present invention;

FIGS. 17A and 17B are graphs illustrating a variation of the thirdaspect of the present invention;

FIGS. 18A and 18B are graphs illustrating an embodiment of a method forcharacterizing a read circuit according to the third aspect of thepresent invention for a first example of charge retention circuit;

FIGS. 19A and 19B are graphs illustrating the embodiment of the readcircuit characterization method for a second example of charge retentioncircuit;

FIG. 20 partially and schematically shows a variation of the readcircuit compatible with the characterization method of FIGS. 18A, 18B,19A, and 19B; and

FIG. 21 shows an embodiment of a charge retention circuit according toan embodiment of the present invention.

DETAILED DESCRIPTION

The same elements have been designated with the same reference numeralsin the different drawings which have been drawn out of scale. Forclarity, only those elements useful to the understanding of the presentinvention have been shown and will be described. In particular, what useis made of the time data obtained by the circuit according to any of theaspects of the present invention has not been detailed, the presentinvention being compatible with any conventional exploitation of suchtime data. Similarly, the methods and elements at the origin of aprogramming or time countdown initialization have not been detailed, thepresent invention being here again compatible with any need for startingof a time countdown.

FIG. 1 is a schematic block diagram illustrating an electronic device 1comprising an electronic charge retention circuit 10 according to any ofthe aspects of the present invention.

Device 1 is any electronic device capable of exploiting datarepresentative of a time elapsed between two events. It is equipped withan electronic charge retention circuit 10 (Δt) controllable for a timemeasurement. Circuit 10 is likely to be submitted to a supply voltageValim applied between two terminals 13 and 12, terminal 12 beingconnected to a reference voltage (for example, the ground). VoltageValim is used to initialize a charge retention phase. Two terminals 14and 15 of circuit 10 are intended to be connected to a measurementcircuit 11 (MES) capable of converting data about a residual charge ofan element of circuit 10 into data relative to the time elapsed betweenthe retention phase initialization time and the measurement time.Terminal 15 may be used as a reference for the measurement and begrounded.

Circuit 10 is preferentially made in the form of an integrated circuitfrom a semiconductor substrate, for example, made of silicon.

FIG. 2 shows the electric diagram of a first embodiment of acontrollable charge retention circuit 10 according to the first aspectof the present invention.

Circuit 10 comprises a first capacitive element C1 having a firstelectrode 21 connected to a floating node F and having its dielectricspacer 23 designed (by its permittivity and/or by its thickness) to haveleakages which are not negligible with time. “Floating node F” is usedto designate a node which is not directly connected to any diffusedregion of the semiconductor substrate and, more specifically, which isseparated by a dielectric spacer, from any voltage-application terminal.By default, second electrode 22 of capacitive element C1 is eitherconnected (dotted lines in FIG. 2) to terminal 12 intended to beconnected to a reference voltage, or left unconnected.

A second capacitive element C2 has a first electrode 31 connected tonode F and a second electrode 32 connected to terminal 12. Dielectricspacer 33 of capacitive element C2 exhibits a charge retentioncapacitance greater than that of capacitive element C1.

Preferably, a third capacitive element C3 has a first electrode 41connected to node F and a second electrode 42 connected to terminal 13of the circuit, intended to be connected to a power source (for example,voltage Valim) on initialization of a charge retention phase.

A function of capacitive element C2 is to store an electric charge. Afunction of capacitive element C1 is to relatively slowly dischargestorage element C2 (as compared with a direct grounding of its electrode31) due to the leakage through its dielectric spacer. The presence ofcapacitive element C2 enables dissociating the charge level present inthe circuit from the discharge element (capacitance C1). The thicknessof the dielectric of element C2 is greater than that of element C1. Thecapacitance of element C2 is greater, preferably by a ratio of at least10, than that of element C1.

A function of capacitive element C3 is to enable an injection of chargesinto capacitive element C2 by Fowler-Nordheim effect or by a hotelectron injection phenomenon. Element C3 enables avoiding the stress onelement C1 on charge of elements C2 and C1 in parallel. The thickness ofthe dielectric spacer of element C3 is greater than that of element C1,to avoid introducing a parasitic leakage path.

Node F is connected to a gate G of a transistor with an isolated controlterminal (for example, a MOS transistor 5), having its conductionterminals (drain D and source S) connected to output terminals 14 and 15to measure the residual charge contained in element C2 (neglecting thecapacitance of element C1 in parallel). For example, terminal 15 isgrounded and terminal 14 is connected to a current source enablingcurrent-to-voltage conversion of drain current I₁₄ in transistor 5.

The thickness of the gate dielectric of transistor 5 is greater thanthat of the dielectric of element C1 to avoid introducing an additionalleakage on node F. Preferably, the gate thickness of transistor 5 iseven greater than the thickness of the dielectric of element C3, toavoid introducing a parasitic programming path (of injection orextraction of charges into and from node F).

FIG. 3 shows an example of the shape of drain current I₁₄ of transistor5 according to voltage V_(F) at node F, referenced with respect toterminal 15. Voltage V_(F) then expresses the gate-source voltage oftransistor 5. It depends on the residual charge across capacitances C1and C2 in parallel, and thus essentially on the residual charge incapacitance C2. The evaluation of drain current I₁₄ may be performed bymaintaining terminals 12 and 15 at the same voltage (for example, theground) and by applying a known voltage on terminal 14. Differentreference voltages may also be applied on terminals 12 and 15, as willbe seen hereafter in relation with FIGS. 13 and 14.

FIG. 4 illustrates the variation of charge Q_(F) at node F along time.At a time t0 when voltage Valim stops being applied on terminal 13,charge Q_(F) starts from an initial value Q_(INIT), to annul at a timet1 with a capacitive discharge shape. The time interval between times t0and t1 not only depends on the leakage capacity of the dielectric ofelement C1, but also on the value (and thus on the storage capacity) ofelement C2, which conditions value Q_(INIT).

Assuming that terminals 12, 15 and second electrode 22 of capacitiveelement C1 are at reference voltages and that terminal 14 is biased to adetermined level so that a variation in current I₁₄ only results from avariation in the voltage of node F, this variation then only depends onthe time elapsed from time t0.

Such a result can be obtained due to the dissociation performed betweenthe time leakage element (C1) and the element representative of theresidual charge (C2).

The programming or resetting of the circuit through capacitive elementC3 protects capacitive element C1 which has a relatively thin oxidethickness (dielectric) and which would otherwise risk being deterioratedin the programming. This especially enables making the measurementsreliable and reproducible along time.

Several capacitive elements C3 may be connected in parallel betweenterminal 13 and node F to accelerate the programming or reset time.

Similarly, the retention time may be adapted not only by setting thethicknesses and/or the permittivities of the dielectrics of elements C1and C2, but also by providing several elements C1 and/or C2 in parallel.

FIG. 5 shows a second embodiment of a circuit according to the presentinvention. As compared with the embodiment of FIG. 2, transistor 5 isreplaced with a transistor 6 with a floating gate FG connected to nodeF. Control gate CG of transistor 6 is connected to a terminal 16 ofcontrol in read mode of the residual charge in the circuit.

FIG. 6 illustrates, in a graph of current I₁₄ versus voltage V₁₆ appliedon the control gate, the operation of the circuit of FIG. 5. It isassumed that the voltage at drain and source terminals 14 and 15 oftransistor 6 is maintained constant by an external read circuit (11,FIG. 1). The voltage drop between the floating gate and terminal 15 thendepends on the electric charge present at node F, on the totalcapacitance between nodes F and 12 (essentially capacitances C1 and C2),and on the voltage applied on control terminal 16 of transistor 6. InFIG. 6, three curves a, b, and c have been illustrated. Curve a showsthe case where node F is totally discharged. Curve b shows the case of apositive charge present on node F (electron extraction). The thresholdof transistor 6 is then lowered. Curve c shows the case of a negativecharge at node F (electron injection), which generates a higherthreshold for the MOS transistor.

According to the applications, charges may be injected into or extractedfrom node F to modify the characteristic of transistor 6 from curve a toone of curves b and c. Once isolated from the programming voltage, theleakage of capacitance C1 provides curve a along time.

The dielectric thickness, between floating gate FG and the channel(active area) of transistor 6, is greater than that of element C1 andpreferentially greater than that of element C3.

FIG. 7 shows the electric diagram of a variation according to whichcharge injection or extraction element C3 is a MOS transistor 7 with afloating gate. In the example of FIG. 7, the circuit has been shown asconnected in a portion of its environment. For example, drain 42 oftransistor 7 is connected to a current source 18 receiving voltage Valimand its source 73 is grounded. Its control gate 74 receives a controlsignal CTRL intended to turn on transistor 7 when charges need to beinjected. Floating gate 41 of transistor 7 is connected to node F. Thedrain (terminal 14) of transistor 6 receives supply voltage Valim andits source is grounded by a current source 19. Voltage V₁₉ acrosscurrent source 19 is representative of the voltage at node F.

The variation of FIG. 7 provides a structure enabling injection ofelectrons on node F by a so-called hot carrier (electrons) phenomenon,by applying adapted voltages between terminals 42, 73, and 74.

After, an electron extraction (application on terminal 13 of a positivereset voltage with respect to terminal 12) by Fowler-Nordheim effect isassumed, but the operation which will be described easily transposes toan injection of electrons at node F, for example, by a so-called hotcarrier phenomenon.

There appears from the foregoing description that it is possible todefine a correlation between the residual charge (with respect to theinitial charge) and the time spent after a circuit reset phase.

Any circuit for reading the voltage of node F may be envisaged. Forexample, the measured value of the current in transistor 5 (or 6) or ofa voltage representative of this current may be converted into timebased on a conversion table or, after digitization, on a conversion lawestablished from a characterization of the circuit. A preferred exampleof a read circuit for interpreting the time discharge will be describedin relation with FIG. 13 to 19B.

Although reference has been made to a single supply voltage Valim,different voltages may be used in programming and reading, provided tohave an exploitable reference between the residual charge and themeasurement.

According to a specific example of embodiment, a charge retentioncircuit according to the first aspect of the present invention is formedwith the following values:

capacitance C1: 2 fF, dielectric thickness: 40 angstroms;

capacitance C2: 20 fF, dielectric thickness: 160 angstroms;

capacitance C3: 1 fF, dielectric thickness: 80 angstroms.

Such a circuit initialized by application of a voltage on the order of12 volts is discharged after approximately one week. This of course isan example only, the values of the dielectric thicknesses, thedielectric constants, and the possible parallel association of severalelements C1 or C2 conditioning the charge retention time.

FIGS. 8A, 8B, 8C, 9A, 9B, 9C, 1A, 10B, 10C, 11A, 11B, 11C, 12A, 12B, and12C show the forming of a circuit according to the embodiment of FIG. 7in an integrated structure derived from an EEPROM memory architecture,according to the second aspect of the present invention.

FIGS. 8A, 9A, 10A, 11A, and 12A are simplified top views, respectivelyof the electronic charge retention circuit and of its elements C2, 7,C1, and 6. FIG. 8B is a cross-section view along line AA′ of FIG. 8A.FIGS. 9B, 10B, 11B, and 12B respectively are cross-section views alonglines BB′ of FIGS. 9A, 10A, 11A, and 12A. FIGS. 8C, 9C, 10C, 11C, and12C show the respective equivalent electric diagrams of the electroniccharge retention circuit and of its elements C2, 7, C1, and 6.

In the described example, an N-channel transistor implementation in aP-type silicon substrate is assumed. The opposite is of course possible.

Each element or cell C2, 7, C1, or 6 is obtained from a floating gatetransistor series-connected with a single-gate selection transistor T2,T3, T1, or T4 for selecting, for example from an EEPROM memory cellarray, the electronic charge retention circuit.

The floating gates of the different transistors forming elements C2, 7,C1, and 6 are interconnected (conductive line 84) to form floating nodeF. Their control gates are connected together to a conductive line 85 ofapplication of read control signal CG. Their respective sources areinterconnected to terminal 12 (the ground) and their respective drainsare connected to the respective sources of selection transistors T2, T3,T1, and T4.

The gates of transistors T1 to T4 are connected together to a conductiveline 86 of application of a circuit selection signal SEL. Theirrespective drains D1 to D4 are connected to individually-controllablebit lines BL1 to BL4. The order of the bit lines in FIG. 8C has beenarbitrarily illustrated as BL2, BL3, BL1, and BL4, but the order of thedifferent elements C2, 7, C1, and 6 in the horizontal row direction (inthe orientation of the drawings) is indifferent.

In this example of embodiment, N-type source and drain regions (FIG.8B), separated from one another in the line direction by insulatingareas 81, are assumed. The floating gates are formed in a firstconductive level M1 separated from the active regions by an insulatinglevel 82 and the control gates are formed in a second conductive levelM2 separated from the first one by a third insulating level 83. Thegates of the selection transistors are formed, for example, in level M1.

A difference with respect to a conventional EPROM memory cell array isthat the floating gates are interconnected by groups of four transistorsto form floating node F. Another difference is that the floating-gatetransistors forming the different circuit elements are different fromone another across the thickness of their tunnel window and/or in theirdrain and source connection.

FIGS. 9A to 9C illustrate the forming of storage capacitor C2. Drain DC2and source SC2 of the corresponding floating-gate transistor areshort-circuited (by extension of the N⁺-type implantation across theentire active area, FIG. 9B) to form electrode 32 of the capacitor.Further, the tunnel window is eliminated with respect to a standardEEPROM cell.

FIGS. 10A to 10C illustrate the forming of transistor 7 formingcapacitive programming element C3. It is a standard EEPROM cell havingthe extension 101 of its N doped area under tunnel window 102 (FIG. 10B)providing a platform in the charge injection area. As a standard EEPROMcell, drain area D7 is connected to the source of selection transistorT3. Source area S7 is connected to terminal 12.

FIGS. 11A, 11B, and 11C illustrate the forming of capacitive element C1forming the leakage element of the charge retention circuit. As comparedwith a standard EEPROM cell, a difference comprises thinning (area 112,FIG. 11B) the dielectric window used for the tunnel effect to increaseleakages. For example, the thickness of dielectric 112 is selected to beapproximately half (for example, between 30 and 40 angstroms) that (forexample, between 70 and 80 angstroms) of a tunnel window (102, FIG. 10B)of an unmodified cell.

FIGS. 12A, 12B, 12C illustrate the forming of read transistor 6, inwhich the tunnel window has been eliminated as well as, preferably, theusual implanted area (101, FIG. 10B) of an EEPROM cell. The active arealimited by source S6 and D6 is thus similar to that of a normal MOStransistor.

The representations of FIGS. 8A to 12C are simplified and may be adaptedto the used technology. In particular, the gates has been shown asaligned with the limits of the drain and source areas, but a slightoverlapping is often present.

An advantage of the forming by means of an EEPROM cell technology isthat the charge retention circuit may be programmed and reset byapplying the same voltage levels and the same time windows as those usedto erase or write into EEPROM memory cells.

Another advantage is that this preserves stability along time whileavoiding degradations of the thin oxide of the leakage element (C1) insuccessive write operations.

The respective connections of bit lines BL1 to BL4 depend on the circuitoperating phases and especially on the programming (reset) or readphase.

Table I hereinafter illustrates an embodiment of a reset (SET) of and areading (READ) from an electronic charge retention circuit such asillustrated in FIGS. 8A to 12C.

TABLE I SEL CG BL2 BL3 BL1 BL4 12 SET VPP₁ 0 HZ VPP₂ HZ HZ HZ READV_(SEL) V_(READ) HZ HZ HZ V₁₄ 0

In a reset phase SET, selection signal SEL is brought to a first highvoltage VPP₁ with respect to ground to turn on the different transistorsT1 to T4 while signal CG applied on the control gates of the floatinggate transistors remains at low level 0 so as to turn on transistor 6.Bit lines BL1, BL2, and BL4 remain floating (high-impedance state HZ)while line BL3 is applied a positive voltage V_(PP2) enabling charge offloating node F. Line 12, common to the sources of the floating-gatetransistors, is preferentially left unconnected HZ.

For the reading READ, the different selection transistors are activatedby signal SEL to a level V_(SEL) and a read voltage V_(READ) is appliedto the control gates of the different floating gate transistors. LinesBL1, BL2, and BL3 are in a high-impedance state HZ while line BL4receives a voltage V₁₄ enabling supply of the read current source. Line12 is here grounded.

The relations between the different levels VPP₁, VPP₂, V_(SEL),V_(READ), and V₁₄ are, preferably, the following:

VPP₁ greater than VPP₂;

V_(SEL) greater than V_(READ);

V_(READ) of the same order of magnitude as V₁₄.

According to a specific example of embodiment:

VPP₁=14 volts;

VPP₂=12 volts;

V_(SEL)=4 volts;

V_(READ)=2 volts; and

V₁₄=1 volt.

What has been described hereabove in relation with one EEPROM cell perelement of the charge retention circuit may of course be replaced with astructure in which subsets of several identical cells in parallel areused for the different respective elements. In particular:

several elements C2 may be used in parallel to increase the capacitanceof node F to increase the electronic circuit discharge time;

several elements 7 may be used in parallel to increase the electroninjection or extraction speed at node F on reset of programming;

several leakage elements C1 may be used in parallel to decrease thesystem discharge time; and/or

several read elements 6 may be introduced in parallel to provide agreater current on evaluation of the circuit.

An electronic retention circuit may be introduced in any position of astandard EEPROM memory cell array, which enables making more difficultits locating by a possible ill-meaning user.

As a variation, several circuits may be placed at different locations ofan EEPROM memory plane. In this case, it may be provided for all thecircuits to have the same discharge time or for the circuits to havedischarge times different from one another.

According to another variation, although several circuits aredistributed in the memory plane, a single one is used at once, accordingto a determined or random sequence controlled by an address generator.

The cell selection transistors forming the charge retention circuit ofthe present invention may be shared with normal EEPROM cells on the samebit lines, provided to provide adapted addressing and switching means.

FIG. 13 shows a first embodiment of a circuit (11, FIG. 1) for readingthe state of an electronic charge retention circuit for a timemeasurement according to the third aspect of the present invention. Forsimplification, the charge retention circuit (FIG. 2, FIG. 5, FIG. 7, orFIGS. 8A to 12C) has been symbolized by a block 10 containing the readtransistor (in this example, a MOS transistor 5) and a capacitiveelement combining elements C1 and C2.

More generally, according to this third aspect of the present invention,the charge retention circuit may be formed of any circuit (for example,that described in above-mentioned international patent applicationWO-A-03/083769).

Output transistor 5 of circuit 10 is placed in a first branch of adifferential assembly comprising two parallel branches of MOStransistors in series between a terminal 131 of application of a supplyvoltage Valim and the ground. Each branch comprises, in series, aP-channel transistor P1 or P2, an N-channel transistor N1 or N2, and anN-channel transistor N3 or N5. The two transistors P1 and P2 have theirgates connected to the source of transistor P2 and their drainsconnected to supply terminal 131. Transistors N1 and N2 have their gatesconnected to a terminal 132 of application of a reference voltage. Thisreference voltage is provided, in this example, by an operationalamplifier 133 receiving on a non-inverting input (+) a voltage V0 andhaving its inverting input (−) connected to the source of transistor N2and to the drain of transistor N5 (terminal 14 of circuit 10). Optionalassembly 133, N1 and N2 enables setting a same voltage level on thesources of transistors N1 and N2. The gate of transistor N3 receives ananalog signal V_(DAC) provided by a digital-to-analog converter 134, theoperation of which will be described hereafter. Its function is toprovide a stepped voltage to interpret the residual charge in circuit10.

The respective sources of transistors P2 and P1 are connected on twoinputs, for example, non-inverting (+) and inverting (−), of acomparator 135 with an output OUT which is used to trigger (TRIGGER 136)the provision of a result TIME corresponding to a binary wordrepresentative of the state COUNT of a counter of the converter. Thiscounter counts at the rate of a clock frequency CK to generate thestepped signal, as will be seen hereafter.

The circuit of FIG. 13 performs a comparison of the difference betweenthe currents in the two branches. The output of comparator 135 switcheswhen the current in branch P1, N1, and N3 becomes greater (or loweraccording to the initial state) than the current in branch P2, N2, andN5.

If terminal 12 is grounded, for a current I₁₄ to flow in the firstbranch, quantity Q_(F)/C_(T) should be greater than the thresholdvoltage (V_(t)) of transistor 5, where Q_(F) represents the residualcharge in circuit 10 and C_(T) represents the cumulated value of thecapacitances between node F and the ground (in particular, capacitiveelements C1 and C2).

Voltage V0 imposed on terminal 14 via amplifier 133 preferablyoriginates from a circuit 137 comprising a follower-assembled amplifier138 (output connected to the inverting input (−)) having itsnon-inverting input (+) connected to the drain of a diode-assembledN-channel transistor N4. The source of transistor N4 is grounded whileits drain is connected, by a constant current source 139 (I0), to aterminal of application of a positive supply voltage (for example,Valim).

Circuit 137 generates a level V0 such that transistor 5 is conductive toenable the reading.

Current I0 is selected according to the consumption wanted for thecircuit.

The N-channel transistors are matched for accuracy reasons.

Preferably, a level greater than level V0 is imposed on terminal 12. Anobjective is, even if cell 10 is entirely discharged, to have transistor5 conduct and enable reading across the entire operating range. Thus,the output of comparator 135 switches when voltage V_(DAC) provided byconverter 134 exceeds level V0+Q_(F)/C_(T).

FIG. 14 shows a preferred embodiment in which a reference structure 10′having its node F’ permanently discharged is used to set the voltage ofterminal 12 of circuit 10. For example, a transistor 140 (Pass Gate)connects terminals 12 and 12′ of circuits 10 and 10′. An amplifier 141has its non-inverting input (+) connected to terminal 14′ of circuit 10′and, by a constant current source 142 (I0), to terminal 131 ofapplication of the supply voltage. The inverting input (−) of amplifier141 receives reference voltage V0 generated by a circuit 137 such asdescribed in relation with FIG. 13. Current sources 139 and 142 generatea same current I0. Accordingly, the voltage of terminal 14′ is set to V0(imposed by the feedback of amplifier 141 and by the gate of transistor5′, which is at level V0, by the sizing of source 142). The voltage ofterminal 12′ is greater than level V0 even if no charge is stored atnode F′. Indeed, when a voltage is applied on terminal 12′ (by amplifier141), node F′ represents the midpoint of a capacitive divider (be itonly by taking into account the gate capacitance of transistor 5′ withrespect to ground). Accordingly, to obtain level V0 at node F′, thevoltage of terminal 12′ is greater than level V0.

To simplify the description of FIG. 14, the rest of the structure,identical to that discussed in relation with FIG. 13, has not beendetailed.

Transistor 140 is only turned on in read mode of the circuit. The restof the time, terminal 12 is either unconnected, or grounded.

When transistor 140 is on, the voltage of terminal 12′ is transferred toterminal 12. Since the voltage of terminal 14 is set to level V0 byamplifier 133 (having its non-inverting input connected to the output ofcircuit 137), the voltage of node F is at level V0 plus the chargestored on this node. If cell 10 is not charged, node F is at level V0.If the cell contains a charge Q_(F), the voltage at node F is equal toV0+Q_(F)/C_(T).

An advantage of this embodiment where transistor 140 sets the samevoltage on the second accessible electrodes of the capacitive elementsof circuits 10 and 10′ is to compensate for possible manufacturingdispersions.

Be it the read circuit of FIG. 13 or of FIG. 14, it can be turned off bymeans of adapted control switches (for example, disconnecting the powersupply branches and/or turning off the current sources) outside of readperiods.

On the read side, assuming that charge Q_(F) has an initial valueQ_(INIT), here noted Q(r), a stepped voltage V_(DAC) provided byconverter 134 ranging between V0 and V0+Q(r)/C_(T) enables measuringtime.

Starting from a level V0+Q(r)/C_(T) and progressively decreasing thelevel, the switching point of comparator 135 corresponds to a digitalreference point COUNT of the converter. This reference point is aninformation as to the time elapsed since the reset (programming ofcharge retention circuit 10) at level Q(r). Examples will be given inrelation with FIGS. 16A to 19B.

An advantage is that the outputting of a digital word is easilyexploitable.

Preferably, the digital-to-analog converter is a non-linear converter tocompensate for the non-linear curve (FIG. 4) followed by the capacitivedischarge of the charge retention circuit. As a variation, thecorrection is performed downstream by digital means (of calculator type)correcting the elapsed time according to count COUNT at which the readcircuit switches.

FIG. 15 shows an example of an electric diagram of a digital-to-analogconverter 134. A reference voltage V_(ref) is provided on a differentialamplifier 151 having its output connected to the common gates of n+2branches comprising a P-channel MOS transistor 152, 152 ₀, 152 ₁, . . ., 152 _(n). A first transistor 152 has its source grounded by a resistorR and connected to the inverting input (−) of amplifier 151 to set aV_(ref)/R current. Transistors 152 ₀ to 152 _(n) of the next n+1branches 152 ₀ to 152 _(n) are of increasing size from one branch to thenext one, starting from the unity size of transistor 152 ₀, equal tothat of transistor 152. The size ratio is preferably double from onebranch to the next one to reflect the binary character of the countingon the voltage amplitudes. The respective sources of transistors 152 and152 ₀ to 152 _(n) are connected to a terminal 150 of application of asupply voltage Valim. The respective drains of transistors 152 ₀ to 152_(n) are connected, by switches K₀ to K_(n), to the drain of a N-channelMOS transistor 155 assembled as a diode and as a current mirror on asecond N-channel transistor 156. The sources of transistors 155 and 156are grounded. The drain of transistor 156 is connected to an invertinginput (−) of an operational amplifier 157 having its non-inverting input(+) receiving reference voltage V0 of the read circuit and having itsoutput providing voltage V_(DAC). A resistor R′ (for example, of samevalue as resistor R) connects the output of amplifier 157 to itsinverting input. Switches K₀ to K_(n) (for example, MOS transistors) arecontrolled by respective bits b0, b1, . . . , bn of a counting circuitover n+1 bits. The counting circuit comprises a counter 153 having n+1bits sent in parallel onto a non-linear conversion circuit 154 (NLC).Amplifiers 151 and 157, as well as counter 153 and circuit 154, aresupplied, for example, with voltage Valim.

Assuming resistors R and R′ to be of the same value, the current intransistor 156 is equal to k*V_(ref)/R, where k represents state COUNTof the counting circuit. Output voltage V_(DAC) is then provided byrelation V0+k*V_(ref).

Other non-linear digital-to-analog conversion circuits may be used, thecircuit of FIG. 15 representing a simple example of embodiment of such aconverter.

FIGS. 16A and 16B illustrate a first operating mode of a read circuitaccording to the third aspect of the present invention and respectivelyshow examples of variations of voltage Q_(F) and of voltage V_(DAC)along time.

An initialization of the discharge circuit at a level Q(r) at a time t0and a reading at a time t_(R) where the residual charge is Q_(R) areassumed.

The non-linearity of the converter is defined by circuit 154 tocompensate for the charge retention circuit discharge curve, forexample, based on experimental or characterization data. Circuit 154 is,for example, a combinational logic converting a linear increase of theoutput of counter 153 into a non-linear increase.

According to the time at which the reading is performed (for example,t_(R), FIG. 16A), the current in transistor 5 generates a switching ofoutput OUT with a delay Δs with respect to the read beginning time (timeorigin of the timing diagram of FIG. 16B). This time interval actuallycorresponds to a number provided by counter 153 in the generation of thestepped voltage sent onto the gate of transistor N3 (FIG. 13). The stateof the counter at the time when signal OUT switches enables deducing thetime interval Δt elapsed between programming time t0 and read timet_(R), whether the device containing the charge retention circuit has ornot been supplied (provided for its terminal 13 to have remainedunconnected or isolated). In the example of FIGS. 16A and 16B, a voltageV_(DAC) decreasing from level V0+Q(r)/C_(T) is assumed. A measurement byincreasing voltage is of course possible, switching point t_(S)remaining the same.

The rate of the steps of voltage V_(DAC) (and thus frequency CK ofcounter 153) is selected to be sufficiently fast with respect to thedischarge speed of circuit 10 for interval Δs between the read beginningtime t_(R) and switching time t_(S) to be negligible with respect toreal interval Δt (t_(R)−t₀). The exaggeration of the representation ofthe drawings however shows the opposite.

It can thus be seen that the discharge of element 10 of the presentinvention may be performed with no power supply, without for all this toloose the time notion.

Voltage V_(ref) is preferably selected to comply with equationk*V_(ref)=Q(r)/C_(T).

Preferably, an adjustment of the read circuit is performed by storing,in a non-volatile memorization register 158 (NVM), a voltage valueV_(ref) or starting number k of the counter obtained by characterizationto comply with the above relation, and by using this value on eachreading.

FIGS. 17A and 17B show, in two initial charge states Q(r′) and Q(r″),examples of decrease in the charge along time and the possibleadjustment performed with the non-linear digital-to-analog converter.

The fact of adjusting the reference value (in this example, respectivelyat values Q(r′)/(k*C_(T)) and Q(r″)/(k*C_(T)) makes the time measurementindependent from the programming conditions, that is, from initialcharge Q(r′) or Q(r″). As can be seen in FIGS. 17A and 17B, switchingtime t_(S) is the same while the converter starting levels aredifferent, as they are adapted to the initial charge levels.

According to whether the discharge curve is known or not, it may benecessary to calibrate each discharge circuit 10 so that thenon-linearity of converter 134 follows the discharge curve.

FIGS. 18A, 18B, 19A, and 19B illustrate a preferred embodiment of thepresent invention in which a calibration of the read circuit isperformed in a first use, in an initialization, or at the end of themanufacturing. For this purpose, the circuit is programmed at a timet10, then measured at a time t11, its interval with respect to time t10being known (for example, a 24-hour interval). The number of steps ofthe stepped decrease provided by the digital-to-analog converter untilswitching time t_(S) is then determined. This enables defining, for theconcerned circuit, the number of steps or stages for the known timeinterval. This number can then be stored in a non-volatile storageelement of device 1.

FIGS. 18A and 18B illustrate a first example in which 7 steps arerequired for 24 h. The time interval (TIME STEP) between two steps isthen 24/7.

FIGS. 19A and 19B illustrate a second example in which 13 steps arerequired to define a same time range by means of another differentcircuit, for example, by the values of capacitances C1 and C2. The timeinterval between two steps then is 24/13.

FIG. 20 is a schematic block diagram partially illustrating an exampleof possible adaptation of the circuit of FIG. 15 to obtain the operationof FIGS. 18A, 18B, 19A, and 19B. This modification comprises using countCOUNT provided by counter 153 to multiply it (multiplier 160) by a timeconversion parameter (Δt/STEP) stored in the non-volatile memory (block161, NVM), to provide a modified counting value COUNT′ taking intoaccount the circuit characteristics. Value COUNT′ is provided to trigger136. This amounts to applying a weighting coefficient which is afunction of an initial circuit characterization measurement.

An advantage of this embodiment is that it requires no structuralmodification of the read circuit to adapt to different charge retentioncircuits.

FIG. 21 is a schematic block diagram illustrating an embodiment of acharge retention circuit in an example of environment implementing thefourth aspect of the present invention.

This drawing is based as an example on the embodiment of the chargeretention circuit shown in FIG. 2. Terminal 13 is connectable by aswitch 211 controlled by programming signal SET to a voltage VPP₂ forinitializing a discharge period. Terminal 14 is connectable, by a switch212 controlled by read signal READ, to a read voltage V₁₄, voltage V₁₉across current source 19 (illustrated by a resistor) providing datarepresentative of the time elapsed since the initialization.

According to a preferred example of the fourth aspect of the presentinvention, element C1 is also usable as a fast programming element byapplying adapted voltage levels to obtain a fast injection or extractionof electrons on node F. A switch 213 then is interposed betweenelectrode 22 of element C1 and a terminal of application of a voltageVPP₃ to force a charge injection or extraction on node F. Switch 213 iscontrolled by a fast programming signal FLASH SET. In the quiescentstate (when it does not apply voltage VPP₃ on electrode 22), switch 213at least functionally grounds electrode 22. In practice, switch 213 mayleave terminal 22 unconnected. It is enough for a discharge path toexist, due to the circuit structure, from node F to the ground throughleakage element C1. Such is in practice almost always the case.

The example described in relation with FIG. 21 is particularly welladapted to a charge retention circuit formed from floating-gatetransistors (FIGS. 8A to 12C).

Such a fast programming (relatively fast as compared with the normalprogramming by element C3) may be used, for example, after a detectionof an abnormal operating condition aiming at preventing the normalcircuit programming.

The risk of stressing the dielectric of element C1 and thus losing thereproducibility of the measurements is acceptable since this case is inprinciple uncommon along the product lifetime. Further, any alterationof the dielectric tends to accelerating the discharge, and thus reducethe time window. Now, such is most often the desired effect in case ofan abnormal operation. In particular, if such an operation is providedin case an attempt for hacking a product is detected, decreasing thecapacity of use on each detection follows the line of generally desiredprotections.

According to the applications, the fast programming function may be usedeither to bring charges onto node F and restart a time period, orconversely to force a fast discharge of node F, for example, to forbid asubsequent access to data protected by the charge retention circuit.

Table II hereafter illustrates an embodiment of a fast programming(FLASH SET) according to the fourth aspect of the present invention inan embodiment of the charge retention circuit of the type illustrated byFIGS. 8A to 12C. Table II shows the programming and read phases ofabove-described table I.

TABLE II EL CG BL2 BL3 BL1 BL4 12 SET PP₁ 0 HZ VPP₂ HZ HZ HZ FLASH SETPP₁ 0 HZ HZ VPP₃ HZ HZ READ SEL V_(READ) HZ HZ HZ V₁₄ 0

Fast programming FLASH SET comprises applying bias voltage VPP₃ (forexample, equal to the level VPP₂ which is available) on line BL1 (FIG.8C), while all the other bit lines BL2 to BL4 are in high impedancestate HZ, and a zero signal CG while signal SEL at level VPP₁ turns onselection transistors T1 to T4. Line 12 preferentially is in ahigh-impedance state HZ.

The fast programming takes advantage of the low dielectric thickness ofelement C1 with respect to dielectric 102 (FIG. 10B) of reset transistor7 to accelerate the programming.

An advantage of this aspect of the present invention is to combine atime measurement after periods of no supply with a fast programmingfunction in charge or discharge mode.

The present invention finds many applications in any system where a timeis desired to be measured on a non-supplied circuit. A specific exampleof embodiment relates to the management of rights of access to data orprograms stored on digital supports. In such an application, a circuitaccording to the present invention may be added to the storage system(memory key or the like) which is not permanently supplied, or be in aseparate circuit and be reset, for example, on a first loading of thedata to be protected.

A second example of application relates to the measurement of timeintervals between any two elements, for example, in applications oftransactional type.

Of course, the present invention is likely to have various alterations,modifications, and improvements which will readily occur to thoseskilled in the art. In particular, the practical implementation of thepresent invention based on the functional indications given hereaboveand on the needs of the application raises no difficulty. For example,the programming may be accessible only once or be resumed at eachpowering-on of the application. Further, especially since it requires nopermanent power supply, the present invention may be implemented incontactless devices (of electromagnetic transponder type) which drawtheir supply from an electromagnetic field in which they are present(generated by a terminal).

Such alterations, modifications, and improvements are intended to bepart of this disclosure, and are intended to be within the spirit andscope of the invention. Accordingly, the foregoing description is by wayof example only and is not intended as limiting. The present inventionis limited only as defined in the following claims and the equivalentsthereto.

1. A circuit for reading from an electronic charge retention element fora time measurement, of the type comprising at least one capacitiveelement with a dielectric exhibiting a leakage and a transistor with anisolated control terminal for reading residual charges, comprising: twoparallel branches between two power supply terminals, each branchcomprising at least one transistor of a first type and one transistor ofa second type, the transistor of the second type of one of the branchesbeing formed by that of the element to be read and the transistor of thesecond type of the other branch receiving, on its control terminal, astepped signal, the respective drains of the transistors of the firsttype being connected to the respective inputs of a comparator having itsoutput providing an indication of the residual voltage level in thecharge retention element.
 2. The circuit of claim 1, wherein each branchcomprises a second transistor of the second type having its controlterminals connected at the output of an amplifier of application of areference voltage, said voltage being also applied on a conductionterminal of the transistor of the charge retention element.
 3. Thecircuit of claim 1, wherein a reference terminal of the capacitiveelement of the charge retention element to be read from is connectableto a reference terminal of a second reference charge retention element.4. The circuit of any of claims 1, wherein said stepped signal isprovided by a digital-to-analog converter of non-linear type.
 5. Amethod for calibrating the read circuit of claim 1, wherein a voltagereference value of the digital-to-analog converter corresponds toQ(r)/(k*C_(T)), where k represents the maximum count of the converter,represents the initial charge of the retention circuit, and C_(T)represents its capacitance.
 6. A method for calibrating the read circuitof claim 1, wherein a weighting coefficient of the count provided by theconverter is deduced from an initial measurement from a known timeinterval.